Radar transmitter circuitry and techniques

ABSTRACT

A radar transmitter includes a digital ramp generator circuit for generating a VCO control signal. The ramp generator includes a digital signal processor and a digital-to-analog converter. In one embodiment, the VCO output signal is up-converted to provide the transmit signal and in another embodiment, the VCO operates over the transmit frequency. Also described is a VCO comprising a DR and a phase shifter. A temperature compensation feature includes detecting the transmit frequency and comparing the DSP output generating the detected frequency to a DSP output stored in association with the detected frequency. Also described is a technique for compensating for non-linear VCO operation in which the DSP output words are adjusted to provide a waveform complementary in shape to the non-linear VCO characteristic. Susceptibility of the radar to interference is reduced by randomly varying at least one parameter of the ramp signal, such as offset interval or voltage range, in at least one ramp signal cycle.

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application claims the benefit of U.S. ProvisionalApplication No. 60/226,160, filed on Aug. 16, 2000 which application ishereby incorporated herein by reference in its entirety.

STATEMENTS REGARDING FEDERALLY SPONSORED RESEARCH

[0002] Not applicable.

FIELD OF THE INVENTION

[0003] This invention relates generally to radar transmitters and moreparticularly to a radar transmitter having a digital ramp signalgenerator and temperature compensation, non-linear VCO compensation, andreduced interference features.

BACKGROUND OF THE INVENTION

[0004] As is known in the art, there are many types of radartransmission techniques, one of which is Frequency Modulated ContinuousWave (FMCW) transmission, in which the frequency of the transmittedsignal increases linearly from a first predetermined frequency to asecond predetermined frequency. FMCW radar has the advantages of highsensitivity, relatively low transmitter power and good range resolution.

[0005] Various circuitry and techniques can be used to generate an FMCWtransmit signal. One technique is to feed a signal voltage having a rampcharacteristic (referred to herein as a

[0006] Various circuitry and techniques can be used to generate an FMCWtransmit signal. One technique is to feed a signal voltage having a rampcharacteristic (referred to herein as a “ramp signal” or ramp voltage”)to a voltage controlled oscillator (VCO) to generate the frequencymodulated transmit signal commonly referred to as a chirp signal.Typically, the ramp signal is generated by an analog circuit which mayinclude timing pulse generation circuits, integrators and amplifiers.Components of such an analog circuit are fixed at the design stage andthus, such a circuit does not afford much, if any versatility.

[0007] Ideally, the frequency of the VCO output signal varies linearlywith respect to the ramp voltage. When there is non-linearity in theramp signal and/or in the operation of the VCO, the frequency of the RFreturn signal can be spread across an RF frequency range or “smeared”,thereby degrading target detection, resolution and range accuracyperformance of the radar system.

[0008] Another technique for generating an FMCW transmit signal is touse direct-digital synthesis (DDS) in which the transmit signal itselfis digitally synthesized. Typical DDS systems include a phaseaccumulator and a digital-to-analog (D/A) converter. However, thetransmit signal rate is limited by the Nyquist theory to less thanone-half of the maximum clock rate of the D/A converter. Otherdisadvantages of DDS systems include complexity and cost plus anincrease in supporting hardware requirements because of limitations inoperating frequency and tuning range of currently available DDSsynthesizers.

[0009] As is also known, some relatively complex radar systems includemultiple transmit and receive circuits (TRCs) each of which operateindependently of one another. When such transmit and receive circuitsare placed in proximity to one another and operate at the same oroverlapping frequencies, the multiple TRCs can interfere with oneanother, preventing the accurate detection of targets. Other problemscan also arise by simultaneous operation of multiple TRCs.

[0010] Radar systems provide several design challenges. As one example,when radar systems operating at the same or overlapping frequencies areused in proximity to one another, the two systems can interfere with oneanother, preventing the accurate detection of targets. For example,circuit performance variations attributable to temperature changes canresult in interference between multiple TRCs. It would, therefore, bedesirable to provide a radar transmitter circuit which permitsadjustment of transmit signal characteristics in a relatively simplemanner. It would also be desirable to provide a radar transmitter whichcompensates for variations in transit signal characteristics caused byvariations in temperature in the environment in which the radartransmitter is disposed. It would be still further desirable to providean FMCW radar system which compensates for non-linear VCO operation. Itwould be still further desirable to provide a technique which allowssimultaneous operation of multiple TRCs in overlapping frequency ranges.It would be still further desirable to provide a system and techniquewhich allows simultaneous operation of multiple FMCW TRCs.

SUMMARY OF THE INVENTION

[0011] In accordance with the present invention, a radar transmitterincludes a DSP, a D/A converter having an input terminal coupled to theoutput terminal of the DSP and an output terminal at which an analogramp signal is provided, and a VCO having an input terminal responsiveto the analog ramp signal and an output terminal at which a frequencymodulated signal is provided.

[0012] With this arrangement, simple and relatively inexpensivecircuitry is used to generate an analog ramp signal for controlling theVCO. Several advantageous features can be readily implemented byappropriate adjustment of the DSP output words, which results inconcomitant adjustment of the chirp signal. These features include VCOand drive circuit temperature compensation, compensation for non-linearVCO operation, and interference reduction techniques. An analogsmoothing circuit may be coupled between the output terminal of the D/Aconverter and the input terminal of the VCO in order to smooth thestepped D/A converter output.

[0013] In one embodiment, the output of the VCO is up-converted toprovide the transmit signal and in another embodiment, the VCO operatesover the transmit signal frequency range, thereby eliminating the needfor the up-converter.

[0014] Also described is a VCO which consists of a dielectric resonatoroscillator (DRO) to generate the chirp signal. The VCO includes anamplifier, a dielectric resonator (DR) for controlling the centerfrequency of the VCO and a phase shifter for providing a frequencytuning capability to the VCO. The phase shifter is a three terminaldevice which has an input terminal coupled to the amplifier, and anoutput terminal connected to the dielectric resonator. The dielectricresonator is connected back to the input of the amplifier to providepositive feedback and thus create an oscillator. In addition, the phaseshifter has a control terminal to control the frequency of the VCO byproviding a phase shift proportional to the control voltage. Thefrequency modulating signal or “ramp signal” is connected to the controlterminal which is responsive to the ramp signal.

[0015] A temperature compensation feature is described including thesteps of generating, from a predetermined sequence of digital words, atransmit signal having a frequency associated with a respective one ofthe sequence of words, and storing each of the digital words inassociation with an expected transmit signal frequency. The actualfrequency of the transmit signal is detected and the digital word usedto generate the detected frequency is compared to the digital wordstored in association with the actual transmit signal frequency. Theresult of the comparison is an error value which is used to adjust eachof the digital words. In one embodiment, the actual transmit signalfrequency is detected with a circuit which is responsive to a narrowband of frequencies and the digital words are adjusted by introducing anoffset equal to the error value. In one embodiment the circuit isprovided as a DRO which is responsive to signals having the transmitsignal frequency.

[0016] According to a method for compensating for non-linear VCOoperation, the VCO is characterized during manufacture by feeding apredetermined sequence of digital words to the D/A converter anddetecting the resulting transmit signal frequency for each word. Thisprocess yields a so-called VCO curve which relates VCO output frequencyto VCO input voltage. A curve having a shape which is complementary tothe shape of the VCO curve is determined and the DSP output words areadjusted to provide the complementary waveform to the VCO. Bycontrolling the VCO with a waveform complementary with respect to itscharacteristic curve, frequency smear caused by such non-linearoperation is reduced.

[0017] A method for reducing interference between radar systems includesthe steps of generating a ramp signal for controlling a VCO and randomlyvarying at least one parameter of the ramp signal. The ramp signalincludes a plurality of cycles, each having an offset portion, a rampportion, and a CW portion. The parameter of the ramp signal may berandomly varied in one or more of the cycles. Illustrative ramp signalparameters which may be randomly varied include, the starting rampsignal voltage, the duration of the offset portion and the voltage rangeof the ramp portion.

BRIEF DESCRIPTION OF THE DRAWINGS

[0018] The foregoing features of this invention, as well as theinvention itself, may be more fully understood from the followingdescription of the drawings in which:

[0019]FIG. 1 is a block diagram of a radar system;

[0020]FIG. 2 is a block diagram of an automotive near object detection(NOD) system including a plurality of radar systems of the type shown inFIG. 1;

[0021]FIG. 3 is a detailed block diagram of a side object detection(SOD) system suitable for use in the NOD system of FIG. 2;

[0022]FIG. 4 is a plot of VCO control signal voltage vs. time whichillustrates VCO control signal waveforms provided by the ramp generatorof FIG. 3 for implementing a temperature compensation feature;

[0023]FIG. 5 is a plot of VCO control signal voltage vs. time whichfurther illustrates VCO control signal waveforms provided by the rampgenerator of FIG. 3 for implementing an interference reduction feature;

[0024]FIG. 5A is plot of VCO control signal voltage vs. time whichillustrates yet another VCO control signal waveform provided by the rampgenerator of FIG. 3 for implementing an alternative interferencereduction feature;

[0025]FIG. 6 is a plot of VCO control signal voltage vs. time whichillustrates a non-linear relationship between a frequency of the VCOoutput signal of FIG. 3 and a voltage of the VCO control signal of FIG.3;

[0026]FIG. 6A is a plot of VCO control signal voltage vs. time whichillustrates a relationship between the VCO control signal and the outputof the DSP of FIG. 3 used to compensate for the non-linear VCOcharacteristic of FIG. 6;

[0027]FIG. 7 is detailed a block diagram of an alternate SOD systemembodiment;

[0028]FIG. 8 is a detailed block diagram of a further alternate SODsystem embodiment;

[0029]FIG. 9 is a block diagram of an illustrative architecture for theDSP of FIGS. 3, 7, and 8;

[0030]FIG. 10 is a diagrammatic view of an exemplary detection zoneprovided by a SOD system disposed on a vehicle;

[0031]FIGS. 11A and 11B are diagrammatic views of alternative detectionzones which can be provided by the SOD system;

[0032]FIG. 13 is a timing chart of a minor cycle detection processincluded in the process of FIG. 12;

[0033]FIG. 14A is a plot of signal return fast Fourier transform (FFT)magnitude vs. FFT frequency bin which illustrates output generated at anintermediate stage of the detection process of FIG. 12;

[0034]FIG. 14B is a plot of FFT derivative magnitude vs. FFT frequencybin which illustrates first and second derivatives of the FFT output ofFIG. 14A computed at another intermediate stage of the detection processof FIG. 12;

[0035]FIG. 15 is a plot of signal return FFT magnitude vs. FFT frequencybin which illustrates an exemplary FFT output resulting from detectionof a geometrically extended target and an illustrative FFT outputresulting from detection of a non-extended target at the same range; and

[0036]FIG. 16 is a plot of signal return FFT magnitude vs. FFT frequencybin which illustrates exemplary FFT outputs associated with detectionsof two proximate targets.

DESCRIPTION OF THE PREFERRED EMBODIMENT

[0037] Referring to FIG. 1, a radar system 10 includes an antennaportion 14, a microwave portion 20 having both a transmitter 22 and areceiver 24, and an electronics portion 28 containing a digital signalprocessor (DSP) 30, a power supply 32, control circuits 34 and a digitalinterface unit (DIU) 36. The transmitter 22 includes a digital rampsignal generator for generating a control signal for a voltagecontrolled oscillator (VCO), as will be described.

[0038] The radar system 10 utilizes radar technology to detect one ormore objects, or targets in the field of view of the system 10 and maybe used in various applications. In the illustrative embodiment, theradar system 10 is a module of an automotive radar system (FIG. 2) and,in particular, is a side object detection (SOD) module or system adaptedfor mounting on an automobile or other vehicle 40 for the purpose ofdetecting objects, including but not limited to other vehicles, trees,signs, pedestrians, and other objects which can be located proximate apath on which the vehicle is located. As will be apparent to those ofordinary skill in the art, the radar system 10 is also suitable for usein many different types of applications including but not limited tomarine applications in which radar system 10 can be disposed on a boat,ship or other sea vessel.

[0039] The transmitter 22 operates as a Frequency Modulated ContinuousWave (FMCW) radar, in which the frequency of the transmitted signallinearly increases from a first predetermined frequency to a secondpredetermined frequency. FMCW radar has the advantages of highsensitivity, relatively low transmitter power and good range resolution.However, it will be appreciated that other types of transmitters may beused.

[0040] Control signals are provided by the vehicle 40 to the radarsystem 10 via a control signal bus 42. The DSP 30 processes thesecontrol signals and radar return signals received by the radar system 10in order to detect objects within the field of view of the radar system,as will be described in conjunction with FIGS. 10-16. The radar system10 provides to the vehicle one or more output signals characterizing anobject within its field of view via an output signal bus 46. Theseoutput signals may include a target detection signal when a targetexceeds the system preset thresholds. The output signals may be coupledto a control unit of the vehicle 40 for various uses such as blind spotand near object detection.

[0041] The antenna assembly 14 includes a receive antenna 16 forreceiving RF signals and a transmit antenna 18 for transmitting RFsignals. The radar system 10 may be characterized as a bistatic radarsystem since it includes separate transmit and receive antennaspositioned proximate one another. The antennas 16, 18 provide multiplebeams at steering angles that are controlled in parallel as to point atransmit and a receive beam in the same direction. Various circuitry forselecting the angle of the respective antennas 16, 18 is suitable,including multi-position transmit and receive antenna switches.

[0042] Referring also to FIG. 2, an illustrative application for theradar system 10 of FIG. 1 is shown in the form of an automotive nearobject detection (NOD) system 100. The NOD system 100 is disposed on avehicle 120 which may be provided for example, as an automotive vehiclesuch as car, motorcycle, or truck, or a marine vehicle such as a boat oran underwater vehicle or as an agricultural vehicle such as a harvester.In this particular embodiment, the NOD system 100 includes aforward-looking sensor (FLS) system 122, an electro optic sensor (EOS)system 124, a plurality of side-looking sensor (SLS) systems 128 orequivalently side object detection (SOD) systems 128 and a plurality ofrear-looking sensor (RLS) systems 130. In the illustrative embodiment,the radar system 10 of FIG. 1 which is shown in greater detail in FIG. 3is a SOD system 128.

[0043] Each of the FLS, EOS, SLS, and RLS systems is coupled to a sensorprocessor 134. In this particular embodiment, the sensor processor 134is shown as a central processor to which each of the FLS, EOS, SLS, andRLS systems is coupled via a bus or other means. It should beappreciated that in an alternate embodiment, one or more of the FLS,EOS, SLS, and RLS systems may include its own processors, such as theDSP 30 of FIG. 1, to perform the processing described below. In thiscase, the NOD system 100 would be provided as a distributed processorsystem.

[0044] Regardless of whether the NOD system 100 includes a single ormultiple processors, the information collected by each of the sensorsystems 122, 124, 128, 130 is shared and the processor 134 (orprocessors in the case of a distributed system) implements a decision orrule tree. The NOD system 100 may be used for a number of functionsincluding but not limited to blind spot detection, lane changedetection, pre-arming of vehicle air bags and to perform a lane stayfunction. For example, the sensor processor 134 may be coupled to theairbag system of the vehicle 132. In response to signals from one ormore of the FLS, EOS, SLS, and RLS systems, the sensor processor maydetermine that it is appropriate to “pre-arm” the airbag of the vehicle.Other examples are also possible.

[0045] The EOS system 124 includes an optical or IR sensor or any othersensor which provides relatively high resolution in the azimuth plane ofthe sensor. The pair of RLS systems 130 can utilize a triangulationscheme to detect objects in the rear portion of the vehicle. The FLSsystem 122 is described in U.S. Pat. No. 5,929,802 entitled AutomotiveForward Looking Sensor Architecture, issued Jul. 27, 1999, assigned tothe assignee of the present invention, and incorporated herein byreference. It should be appreciated that each of the SLS and RLS sensorsmay be provided having the same antenna system.

[0046] Each of the sensor systems is disposed on the vehicle 120 suchthat a plurality of coverage zones exist around the vehicle. Thus, thevehicle is enclosed in a cocoon-like web or wrap of sensor zones. Withthe particular configuration shown in FIG. 2, four coverage zones 68a-68 d are used. Each of the coverage zones 68 a-68 d utilizes one ormore RF detection systems. The RF detection system utilizes an antennasystem which provides multiple beams in each of the coverage zones 68a-68 d. In this manner, the particular direction from which anotherobject approaches the vehicle or vice-versa can be found. One particularantenna which can be used is described in U.S. patent applicationentitled Slot Antenna for an Array Antenna, filed on Aug. 16, 2001, andassigned application Ser. No. 09/______, and U.S. patent applicationentitled Switched Beam Antenna Architecture, filed on Aug. 16, 2001, andassigned application Ser. No. 09/______, each of which are assigned tothe assignee of the present invention and incorporated herein byreference.

[0047] It should be appreciated that the SLS, RLS, and the FLS systemsmay be removably deployed on the vehicle. That is, in some embodimentsthe SLS, RLS, and FLS sensors may be disposed external to the body ofthe vehicle (i.e. on an exposed surface of the vehicle body), while inother systems the SLS, RLS, and FLS systems may be embedded into bumpersor other portions of vehicle (e.g. doors, panels, quarter panels,vehicle front ends, and vehicle rear ends). It is also possible toprovide a system which is both mounted inside the vehicle (e.g., in thebumper or other location) and which is also removable. The system formounting can be of a type described in U.S. patent application entitledSystem and Technique for Mounting a Radar System on a Vehicle, filed onAug. 16, 2001, and assigned application Ser. No. 09/______, and U.S.patent application entitled Portable Object Detection System, filed onAug. 16, 2001, and assigned application Ser. No. 09/______, each ofwhich are assigned to the assignee of the present invention andincorporated herein by reference.

[0048] Referring also to FIG. 3, the radar system 10 of FIG. 1 for useas a SOD system 128 (FIG. 2) is shown in greater detail. In generaloverview of the operation of the transmitter 22, the FMCW radartransmits a signal 50 having a frequency which changes in apredetermined manner over time. The transmit signal 50 is generallyprovided by feeding a VCO control or ramp signal 86 to a voltagecontrolled oscillator (VCO) 92. In response to the ramp signal 86, theVCO 92 generates a chirp signal 88.

[0049] A measure of transmit time of the RF signal can be determined bycomparing the frequency of a received signal 54 with the frequency of asample 58 of the transmit signal. The range determination is thusprovided by measuring the beat frequency between the frequencies of thesample 58 of the transmit signal and the return signal 54, with the beatfrequency being equal to the slope of the ramp signal 86 multiplied bythe time delay of the return signal. The measured frequency furthercontains the Doppler frequency due to the relative velocity between thetarget and the radar system. In order to permit the two contributions tothe measured frequency shift to be separated and identified, thetime-varying frequency of the transmit signal 50 is achieved byproviding a control signal 86 to the VCO 92 in the form of a linear rampsignal followed by either a CW signal or a ramp with the opposite slope.

[0050] According to one aspect of the invention, the VCO control signal86 is generated with digital circuitry and techniques. In a preferredembodiment, the ramp signal 86 is generated by the DSP 30 and adigital-to-analog converter (DAC) 70. Use of the DSP 30 and DAC 70 togenerate the ramp signal 86 is possible in the SOD system 10 since, ithas been determined in accordance with the present invention, that byproper selection of the detection zone characteristics including but notlimited to detection zone size, shape and resolution, precise linearityof the chirp signal 88 is not necessary. This is a result of thecorrelation that exists between the nonlinearities in the transmit andreceive waveforms at close ranges. With this arrangement, the frequencyof the transmit signal 50 is accurately and easily controllable whichfacilitates implementation of several advantageous and further inventivefeatures. As one example, one or more characteristics of successiveramps in the ramp signal 86 are randomly varied in order to reduceinterference between similar, proximate radar systems, as illustrated byFIGS. 5 and 5A. As another example, temperature compensation isimplemented by appropriately adjusting the ramp signal 86, as will bedescribed in conjunction with FIG. 4. Yet another example iscompensation for non-linearity in the VCO operation, as illustrated byFIGS. 6 and 6A. Further, changes to the SOD system 10 which wouldotherwise require hardware changes or adjustments can be made easily, bysimply downloading software to the DSP 30. For example, the frequencyband of operation of the SOD system 10 can be readily varied, as may bedesirable when the SOD is used in different countries with differentoperating frequency requirements.

[0051] The electronics portion 28 of the SOD system 10 in FIG. 3includes the DSP 30, the power supply 32, and a connector 154 throughwhich signal buses 42, 46 (FIG. 1) are coupled between the SOD system 10and the vehicle 40 (FIG. 1). The digital interface 36 is provided in theform of a controller area network (CAN) transceiver (XCVR) 150 in FIG. 3which is coupled to the DSP 30 via a CAN microcontroller 80. The CANcontroller 80 has a system clock coupled thereto to provide frequencystability. In one embodiment, the system clock is provided as a crystalcontrolled oscillator. An analog-to-digital (A/D) converter 68 receivesthe output of a video amplifier 64 and converts the signal to digitalform for coupling to the DSP 30 for detection processing. In oneembodiment, the A/D converter 68 is provided as a twelve bit A/Dconverter. Those of ordinary skill in the art will appreciate, however,that any A/D converter having sufficient resolution for the particularapplication may be used. A digital signal bus 158 is coupled to antennaswitch driver circuits 103 which in turn control microwave switches 99,101 in order to provide control signals to drive the microwave switcheswhich in turn control antenna steering. Also provided in the electronicsportion 28 of the SOD system 10 is a memory 156 in which softwareinstructions, or code and data are stored. In the illustrativeembodiment of FIGS. 3 and 7, the memory is provided as a flash memory156.

[0052] The DSP 30 provides the digital ramp output signals, or words tothe DAC 70 which converts the ramp words into respective analog signals.An analog smoothing circuit 76 is coupled to the output of the DAC 70 inorder to smooth the stepped DAC output to provide the ramp controlsignal 86 to the VCO 92. The DSP 30 includes a volatile memory device304 (FIG. 9) in which is stored a look-up table containing a set of DSPoutput signals, or words in association with the frequency of thetransmit signal 50 generated by the respective DSP output signal. Thisdata is transferred to data RAM 304 from Flash memory 156 during initialboot up of the system. This data may be corrected from time to time as aresult of temperature effects as described herein. In the illustrativeembodiment, the VCO 92 is an SMV2488 device provided by ZCommunications, Inc. of San Diego, Calif. and the VCO output signal 88has a frequency in the range of 2.31 to 2.54 GHz. The SOD embodiments ofFIGS. 7 and 8 illustrate alternative VCO arrangements for generating thetransmit signal.

[0053] An up-converter circuit 90 up-converts the VCO output signal 88to a higher frequency as is desired for transmission in the illustrativeautomotive NOD system 100. In particular, the signal 88 is up-convertedto a frequency of between 24.01 to 24.24 GHz. The up-converter 90includes a 50 ohm load 136, an amplifier 138, a dielectric resonator(DR) 140, and a mixer 142. The amplifier 138, the dielectric resonator(DR) and the transmission lines 144, 146 form an oscillator circuit inwhich the DR 140 couples energy at its fundamental frequency and withinits passband from transmission line 144 to transmission line 146 inorder to generate an oscillator signal for coupling to mixer 142. In theillustrative embodiment, the oscillator signal on transmission line 144has a nominal frequency of 21.7 GHz. The output of the mixer 142 isfiltered by a bandpass filter 96 and is amplified by an amplifier 94. Aportion of the output signal from amplifier 94, is coupled via coupler95 to provide the transmit signal 50 for further amplification byamplifier 78 and transmission by transmitter antenna 18. Another portionof the output signal from amplifier 94 corresponds to a local oscillator(LO) signal 58 fed to an LO input port of a mixer 60 in the receivesignal path.

[0054] The switch circuits 99, 101 are coupled to the transmit andreceive antenna 16, 18 through a Butler matrix. The antennas 18, 16 andswitch circuits 99, 101, and Butler matrix can be of the type describedin U.S. patent application entitled Switched Beam Antenna Architecture,filed on Aug. 16, 2001, and assigned application Ser. No. 09/______,assigned to the assignee of the present invention, and incorporatedherein by reference. Suffice it here to say that the switch circuits 99,101 and Butler matrix operate to provide the antenna having a switchedantenna beam with antenna beam characteristics which enhance the abilityof the SOD system 10 to detect targets.

[0055] The received signal 54 is processed by an RF low noise amplifier(LNA) 52, a bandpass filter 56, and another LNA 62, as shown. The outputsignal of the RF amplifier 62 is down-converted by a mixer 60, whichreceives a local oscillator signal 58, coupled from the transmitter, asshown. Illustrative frequencies for the RF signals from the amplifier 62and the local oscillator signal 58 are on the order of 24 GHz. Althoughthe illustrated receiver 24 is a direct conversion, homodyne receiver,other receiver topologies may be used in the SOD radar system 10.

[0056] A video amplifier 64 amplifies and filters the down-convertedsignals, which, in the illustrative embodiment have a frequency between1 KHz and 40 KHz. The video amplifier 64 may incorporate features,including temperature compensation, filtering of leakage signals, andsensitivity control based on frequency, as described in a co-pendingU.S. patent application entitled Video Amplifier for Radar Receiver, andassigned application Ser. No. 09/______, filed on Aug. 16, 2001,assigned to the assignee of the present invention, and incorporatedherein by reference.

[0057] The A/D converter 68 converts the analog output of the videoamplifier 64 into digital signal samples for further processing. Inparticular, the digital signal samples are processed by a fast Fouriertransform (FFT) within the DSP 30 in order to determine the content ofthe return signal within various frequency ranges (i.e., frequencybins). The FFT outputs serve as data for the rest of the signalprocessor 30 in which one or more algorithms are implemented to detectobjects within the field of view, as will be described in conjunctionwith FIGS. 10-16.

[0058] The radar system 10 includes a temperature compensation featurewith which temperature induced variations in the frequency of thetransmit signal are compensated by adjusting the ramp signal 86accordingly. For this purpose, the transmitter 22 includes a dielectricresonator (DR) 72 coupled to a microwave signal detector 74. The outputof the microwave detector 74 is coupled to an analog-to-digitalconverter which is included in the CAN controller 80 (FIG. 9) forprocessing by the DSP 30.

[0059] In operation, the DR 72 couples energy to the detector 74 onlywhen the transmit signal 50 has a frequency within a range around thefundamental resonant frequency of the DR (i.e., within the passband ofthe DR). In the illustrative embodiment, the DR 72 has a fundamentalfrequency within the transmit frequency range and a passband which isrelatively narrow compared with the transmit frequency range. In thisillustrative embodiment the DR is provided having a passband on theorder of 10 MHz centered in the middle of the band of operation of thesystem in order to provide sufficient frequency detection resolution aswill become apparent. The detector 74 detects output power from the DR72 and provides an electrical signal indicative of a predetermined levelof output power from the DR.

[0060] Detection of output power from the DR 72 indicates transmissionof the DR's fundamental predetermined frequency. Further, transmissionof the predetermined frequency corresponds to a predetermined DSP outputword which, in turn, corresponds to “an expected” transmission frequencyas specified in the look-up table.

[0061] In operation, when the DSP 30 receives an output signal via theCAN controller 80 from the frequency detector 72 and 74 indicatingtransmission of the predetermined frequency, the expected frequencyassociated with the DSP output is compared to the predeterminedfrequency by a software comparator 82 within the DSP. The measured timeof detection and the commanded frequency is correlated in order to makean accurate frequency measurement. Any discrepancy between the expectedfrequency and the measured frequency indicates that an adjustment to theDSP output is necessary. Stated differently, a difference between thetwo frequencies indicates that the look-up table data needs to becorrected, since the expected frequency is not being transmitted inresponse to the corresponding DSP output.

[0062] By way of a simple example, consider the case where the look-uptable indicates that a DSP output of 11110000 corresponds to a transmitfrequency of 24.20 GHz and the DR 72 has a fundamental frequency of24.20 GHz. Thus, detection of output power from the DR 72 indicates that24.20 GHz is being transmitted. However, assume further that thedetection occurs when the DSP output is given by 11110001. This scenarioindicates that the DSP output words need to be adjusted and,specifically, need to be shifted down by one, in order to transmit thedesired, expected frequency.

[0063] Various techniques may be used to compensate for the errorbetween the transmitted frequency and the expected frequency. As oneexample, an offset equal to the amount of the error (i.e., thedifference between the transmitted frequency and the DSP output wordthat is stored in the look-up table in association with the DRO'sfundamental frequency) may be added or subtracted, depending on thedirection of the offset, each time a new DSP output word is provided anduntil a new error is detected. Stated differently, each subsequent DSPoutput is simply shifted by the amount of the detected error. Sincetemperature generally does not change quickly, this error detection andcorrection may be performed relatively infrequently, such as once every50 ms or 100 ms. It will be appreciated by those of ordinary skill inthe art that other structures and techniques may be used for detectingthe frequency of the transmit signal 50 to feedback to the DSP 30 forthe purpose of adjusting the DSP output words in order to thereby adjustthe frequency of the transmit signal.

[0064] Referring also to FIG. 4, a single cycle of an illustrative rampsignal 220 as provided at the output of the DAC 70 (FIG. 3) is shown. Inoperation according to the above-described temperature compensationfeature, detection of an error between the predetermined transmit signalfrequency (i.e., the DRO fundamental frequency) and the expectedtransmit frequency associated with the DSP output word generating thepredetermined frequency results in modification of the DSP output wordsas described above. The effect of shifting the DSP output words up ordown is a shift of the ramp signal up or down, accordingly. For example,following correction of the DSP output, the nominal ramp signal 220 isshifted down in voltage to provide temperature compensated ramp signal224, as shown.

[0065] An interference reduction feature of the SOD 10 according to afurther aspect of the invention is implemented with a random, orpseudo-random number generator 84, (FIG. 3), as may be provided bysoftware within the DSP 30. The random number generator 84 is used torandomly vary at least one aspect, and parameter of the ramp signal 86.Examples of such parameters are the offset interval of each chirp cycleas illustrated in FIG. 5 and the voltage (and thus frequency) rangegenerated in each chirp cycle as illustrated in FIG. 5A.

[0066] Referring to FIG. 5, a plurality of cycles of an illustrativeramp signal 230, corresponding to ramp signal 86 (FIG. 3), for couplingto the VCO 92 are shown. Each ramp cycle starts at a time T1, T2, T3, .. . and has a ramp portion R1, R2, R3, . . . commencing at an offsettime t1, t2, t3, . . . following the respective cycle start time T1, T2,T3, . . . , as shown. According to the invention, the duration of theoffset interval of each cycle (i.e., intervals t1-T1, t2-T2, t3-T3, . .. ) is randomly selected by the DSP 30. This is achieved by introducinga random delay corresponding to the offset interval into a recursiveprocess by which the ramp signal is generated. In the illustrativeembodiment, each ramp cycle is on the order of 1.1 msec and the offsetinterval of each cycle is randomly selected from one of sixteen possibleintervals between 0 and 0.1 msec. As a result, at any given time, thefrequency of the resulting transmit signal 50 will be different than thefrequency of transmit signals from like radar systems, although thetransmission frequency range of the radars will be the same.

[0067] With this arrangement, two identical SOD radar systems 10operating proximate to each other over the same frequency range will notinterfere with one another since, at any given time, the frequency ofthe transmit signals provided by the systems will vary randomly withrespect to each other. This feature advantageously permits two or moreidentical SODs to be used on a vehicle, as is desirable to reduce theparts count of the NOD system and simplify replacement of one or moremodules within the NOD system. A less desirable approach to reduceinterference between proximate SODs would be to manually manipulate orprogram each SOD to ensure different transmit signals.

[0068] An alternative technique for reducing interference illustrated bythe ramp signal 234 of FIG. 5A entails randomly varying the frequencyrange of the transmit signal 50 during each ramp cycle while stillstaying within the specified frequency range for the SOD system 10. Thisis achieved by introducing a random offset voltage to the ramp signalduring each cycle while keeping the peak-to-peak ramp signal voltageconstant (i.e., moving the ramp up or down while keeping peak-to-peakramp voltage constant). For example, during the cycle commencing at timeT1, the ramp R1 increases from 1 volt to 2.5 volts, thus correspondingto a first frequency range. During the next cycle, the ramp R2 increasesfrom 1.25 volts to 2.75 volts, corresponding to a second differentfrequency range. In the illustrative embodiment, the frequency range ofthe transmit signal 50 is randomly selected from twenty-four differentfrequency ranges.

[0069] Various techniques in either the analog or digital portions ofthe circuitry, are possible for introducing an offset voltage to theramp signal in order to randomly vary the voltage range. As one example,the offset voltage is introduced by the smoothing circuit 76 (FIG. 3).

[0070] It will be appreciated that although the ramps R1, R2, R3, . . .of the ramp signal 234 of FIG. 5A have random offset intervals (i.e.,offset intervals computed as t1-T1, t2-T2, t3-T3, . . . ) as describedabove in conjunction with FIG. 5, the transmit frequency range can berandomly varied without also varying the offset interval. Stateddifferently, the technique of randomly varying the offset interval ofeach ramp cycle and randomly varying the frequency range of each rampcycle can be implemented separately or in combination. It will also beappreciated that although the offset interval of each ramp cycle israndomly varied in FIGS. 5 and 5A and the voltage range is randomlyvaried in each ramp cycle in FIG. 5A, the selected parameter(s) may berandomly varied in one or more of the ramp cycles and need not berandomly varied in every ramp cycle.

[0071] Another feature of the SOD system 10 is compensation fornon-linear operation of the VCO 92 (FIG. 3). FIG. 6 shows a curve 238representing the relationship between the frequency of the VCO outputsignal 88 (FIG. 3) versus the VCO control signal voltage 86, (FIG. 3).The illustrative curve 238 has a typical frequency range on the order of180 MHz and a typical voltage range on the order of 1.25 volts. Althoughin an ideal VCO the relationship is linear as illustrated by curve 236,typically there is some non-linearity, as shown in the curve 238. Thisnon-linearity is compensated, or nulled by adjusting the DSP outputwords accordingly. More particularly, the DSP output words are adjustedin order to provide ramp voltage (i.e., DAC output voltage) that issubstantially complementary with respect to the actual curve 238. Such acomplementary curve 240 is shown in FIG. 6A.

[0072] The operation of VCO 92 is characterized (i.e., a curve like FIG.6 is generated) by feeding a sequence of DSP output words to the DAC 70and measuring the transmit frequency with a test receiver (not shown).Such VCO characterization can be done at manufacture or prior to orafter insertion of the VCO in the SOD system 10 or prior to or afterplacing a SOD system on a vehicle. Having characterized the VCO, thecomplementary curve of FIG. 6A is simply determined by taking thecomplement of the curve with respect to the ideal characteristic(labeled 236 in FIG. 6). A sequence of DSP output words necessary togenerate the VCO control signal 240 of FIG. 6A can then be determined.By way of a simple example, where prior to characterization of the VCO,a DSP output sequence of 00000000, 00000001, 00000010, 00000011, . . .might be used to generate a linear ramp, in order to compensate for theVCO non-linearity, the sequence might be adjusted to 00000000, 00000000,00000001, 00000010, . . . The process of determining the necessarysequence of DSP output words to generate the control signal 240 can beperformed manually or can be automated and preferably is repeatable fromsystem to system. Typically, the correction to the DSP output words ismade in software and stored in memory.

[0073] With knowledge of the way in which VCO output frequency varieswith input voltage, the VCO control voltage 86 can be adjusted in orderto force the output frequency of the VCO to be swept in a desiredmanner. Although the relationship between VCO output and VCO input isnon-linear, the relationship between VCO output and DSP output issubstantially linear.

[0074] Also, it will be appreciated that, given knowledge of the FIG. 6characteristic curve 238 of the VCO 92, the system may be optimized tooperate over a particular input voltage range to the VCO. Specifically,the most linear region of operation of the VCO (as labeled 242 in FIG.6) can be selected as the DAC output voltage range simply by selecting arange of DSP output words necessary to generate the desired DAC outputvoltage range.

[0075] Referring also to FIG. 7, an alternate SOD system 250 differsfrom the SOD system 10 of FIG. 3 in the VCO portion of the transmitter22 which processes the ramp signal 86 to generate the transmit signal50. Portions of the SOD system 250 which are identical to the SOD system10 of FIG. 3 have like reference numbers. Like the VCO 92 of FIG. 3, theVCO 254 is responsive to the ramp signal 86 and provides an outputsignal to a bandpass filter 96 which, in turn, provides an output signalhaving the desired transmit frequency in the range of 24.01 to 24.24GHz.

[0076] The SOD system 250 includes a VCO 254 in the form of a voltagecontrolled DRO 254. The VCO 254 further includes a 50 ohm load 258coupled to an amplifier 270 which, in turn, is coupled to a diode phaseshifter 262. The phase shifter 262, signal coupling paths 268, 272,amplifier 270 and dielectric resonator 266, form an oscillation loop.

[0077] In operation, the DR 266 transfers signals within its passbandfrom transmission line 268 to transmission line 272. The signal isamplified by amplifier 270 until the amplifier saturates. The passbandof the DR 266 is selected to cover the desired frequency range oftransmission which, in the illustrative embodiment is a range of24.01-24.24 GHz. The final frequency of the signal on transmission line268 is controlled by adjusting the phase shift introduced by the phaseshifter 262. In this way, the diode phase shifter 262 provides phasereinforcement on the oscillation loop so that the oscillation loop seeksout the frequency determined by the feedback loop which includes thephase shifter and the DR. In general, a feedback loop will oscillate ata frequency whereby a feedback signal, for example a signal on signalpath 272, achieves zero degrees phase (or any multiple of three hundredsixty degree phase) as it travels around the loop back to its startingpoint, with a loop gain, or gain around the loop, greater than one.Thus, by altering phase shifter 262, the frequency at which the feedbacksignal will achieve zero degrees will be similarly altered, and, so longas the loop gain remains greater than one, the loop will oscillate atthe altered frequency. With the arrangement of FIG. 7, up converting ofthe signal provided by the VCO 254 is not necessary since the VCOoperates within the desired transmit frequency range to generate thetransmit signal 50.

[0078] Referring also to FIG. 8, another alternate SOD system 280differs from the SOD system 10 of FIG. 3 in the VCO portion of thetransmitter and the antenna arrangement. Portions of the SOD system 280which are identical to the SOD system 10 of FIG. 3 have like referencenumbers.

[0079] The SOD 280 includes a VCO 284 receiving ramp signal 86 from theanalog smoothing circuit 76. The VCO 284 operates in the transmitfrequency range of between 24.01 to 24.24 GHz and provides an outputsignal to bandpass filter 96, as shown. Since the VCO 284 operates inthe transmit frequency range, the need for up-converting of the VCOoutput signal (FIG. 3) is eliminated.

[0080] It will be appreciated by those of ordinary skill in the art thatfeatures, components, elements and other portions of the SOD system 10of FIG. 3, the SOD system 250 of FIG. 7, and the SOD system 280 of FIG.8 may be mixed and matched. As one example, it will be appreciated thatthe antenna arrangement of FIG. 3 may be used with the VCO 254 of FIG. 7or the VCO 284 of FIG. 8.

[0081] Referring to FIG. 9, a block diagram of the SOD electronics 28(FIGS. 1, 3, 7, and 8) is shown in greater detail to include the DSP 30,the CAN microprocessor 80, the CAN transceiver 150, power supply 32, andthe memory 156. The memory 156 provides non-volatile storage of data andprogram information. Also shown is the video amplifier 64 providing anoutput signal to the A/D converter 68 which is coupled to port 312 ofthe DSP 30. In one embodiment, the memory 156 is provided as a 128 k×8flash memory and the port 312 is provided as a serial port of the DSP30. The D/A converter 70 receives the digitized ramp signal in the formof a sequence of digital words from the DSP through the port 312, asshown, and is further coupled to the smoothing circuit 76.

[0082] The DSP 30 includes RAM 304 in which data is stored, such as thelook-up table of VCO control signal voltage 86 versus transmitfrequency, and a program RAM 306 in which process instructions arestored, such as software code used to implement the detection algorithmdescribed below. Program and data information stored in memory 156 aretransferred to program RAM 306 and data RAM 304, respectively, uponapplication of power. The illustrative DSP 30 is an ADSP2186Mmicroprocessor manufactured by Analog Devices, Inc. and includes anoscillator 308 operating at 33 MHz which is doubled internally to 66 MHzwhich is the clock rate of the DSP.

[0083] The DSP 30 further includes digital input/output (I/O) port 314at which the antenna control signals are provided on bus 158. The DSP 30communicates with the CAN microprocessor 80 via the digital I/O 314 anda serial port 316 which is further coupled to a serial EEPROM 340. Theserial EEPROM 340 is used to hold data such as calibration constants anddiagnostic test results and trouble codes.

[0084] In the illustrative embodiment, the random number generator 84(FIG. 3) used to implement temperature compensation and the comparator82 (FIG. 3) used to compensate for non-linear VCO operation areimplemented by the DSP as a series of software instructions stored inthe program RAM 306 and executed by the microprocessor. It will beappreciated by those of ordinary skill in the art however that thesefunctional blocks as well as others may be implemented in hardware,firmware, software or a combination of hardware, firmware or software.

[0085] The illustrative CAN microprocessor 80 is a TMS470R1F316, amember of the TMS470 family from Texas Instruments and includes RAM 320,flash memory 322, an A/D converter 328, serial ports 330, digital I/Oports 332, a CAN interface 334, and an oscillator 324, here operating at4.915 MHz. The CAN microprocessor 80 is coupled to the DSP 30 throughserial port 330 and digital I/O 332 and is coupled to the CANtransceiver 150 through digital I/O 332 and the interface 334, as shown.

[0086] The CAN A/D converter 328 has a plurality of multiplexed inputsadapted for receiving various monitoring signals. As examples, theoutput of frequency detector 74 (FIG. 3) is coupled to the CAN A/Dconverter 328. Another optional input to the CAN A/D converter 328 isprovided by a temperature sensor 98 (FIG. 3). The temperature sensorprovides an output signal indicative of the temperature of the receiver22 and may be used separately or in conjunction with features of theoscillator 90 (FIG. 3) to compensate for temperature induced frequencyvariations in the RF VCO 90. Optional detectors for detecting powersupply faults may also be coupled to the A/D converter 328. The CAN A/Dconverter 328 converts the received signals to digital form for furtherprocessing, generally by the DSP 30.

[0087] Referring now to FIG. 10, an exemplary detection zone 500, shownin top view, is an azmuthal region in which a SOD system 504 (alsoreferred to as a SOD sensor 504) which may be of the type describedabove in FIGS. 1-9 is specified to detect objects. Only objects within aspecific detection zone 500 in proximity to an automobile 508 on whichthe SOD system 504 is mounted are included in a detection, and objectsoutside of the detection zone 500 are excluded.

[0088] The detection zone shape, of which the indicated zone 500 is onlyone example, depends upon the application in which the radar system isused. For example, an automobile SOD system is designed to detectobjects in the adjacent lane of traffic including objects within theblind spot of the vehicle. Detection of objects in lanes of trafficbeyond the adjacent traffic lane and outside of the blind would beundesirable. As another example, an automobile FLS system 122 (FIG. 2)used for purposes including but not limited to collision avoidance isdesigned to detect objects generally in front of the vehicle. Thus, eachapplication requires a different shape of detection zone 500.

[0089] The sideward detection zone 500 includes a maximum detection zoneboundary 512, a minimum detection zone boundary 520, and a nominaldetection zone boundary 516 between the maximum and minimum detectionzone boundaries. In a preferred embodiment, the SOD system 504 does notdetect objects outside of the maximum detection zone boundary 512 andalways detects objects within the minimum detection zone boundary 520. Aprobability region 524 exists between the maximum detection zoneboundary 512 and the minimum detection zone boundary 520. The SOD systemmay or may not detect an object in the region 524. Probability region524 exists due to imperfections and tolerances of practical circuitcomponents which make up the SOD system 524. Ideally, the region 524would have a width of 0 meters. Thus, the probability of detection inthe probability zone is between zero and one hundred percent. Anexemplary probability region 524 has a maximum width of 0.6 meters. Thedetection zone 500 further includes a minimum range 528 which is thedistance from the SOD system to the closest object that will bedetected. In the illustrative SOD system 504, the nominal detection zoneboundary has an average width on the order of ten meters and an averagelength on the order of four meters. The minimum range 528 is on theorder of 0.25 meters.

[0090] The size and shape specifications of the detection zone 500 andprobability region 524 dictate many design parameters associated withthe SOD system architecture. For example, in order to provide a SODsystem performance that can transition from no detections to certaindetections within a given probability region 524, the SOD system 504 candetermine the range to an object with a measurement accuracy which isrelatively high compared with the width of the probability region 524.For example, in the case where the probability region has a width of 0.6meters, the measurement accuracy of the SOD system should be on theorder of 0.06 meters.

[0091] As another example of a system parameter determined by thedetection zone 500, the specification of a wide detection zone boundary512-520 in azimuth relative to the SOD system 504, requires a wide radarcoverage in azimuth. Furthermore, a detection zone boundary 512-520 witha complex contour requires that multiple beams be used in the detectionalgorithm, each with a different detection range. The beams related tothe detection zone will be shown in greater detail in association withFIGS. 11A and 11B.

[0092] For another example, the specified minimum range 528 of thedetection zone 500, can preferably be achieved with a systemarchitecture that uses a chirp signal, like the FMCW chirp signaldescribed above in conjunction with FIG. 3. In particular, the FederalCommunications Commission (FCC) has specified that low power radar, atpower of six decibels (6 dB) effective isotropic radiated power (6dB_(eirp)) such as that used by the SOD systems discussed above, canhave a frequency bandwidth no greater than 200 MHz. As will becomeapparent from the following description, the short range requirement 528and the FCC bandwidth requirement together preclude the use ofconventional pulsed radar.

[0093] The FCC 200 MHz bandwidth requirement is met by the SOD systembeing an FMCW system since a slowly varying FMCW signal can be generatedwith a narrow bandwidth of 200 MHz. For example, the SOD system 504generates a transmit signal having a frequency which increases fromapproximately 24.000 GHz to 24.200 GHz in approximately 1 ms and has abandwidth of 200 MHz. In contrast, conventional pulsed radar in a shortrange application cannot meet the 200 MHz FCC bandwidth requirement. Apulsed radar is also limited in its ability to detect objects at shortranges. This is because, in order to operate at the short rangesrequired of the SOD system, a pulsed radar system would require a veryshort radar pulse. Not only is such a short pulse technically difficultto achieve, but also, as the pulse width becomes narrower, the bandwidthof the transmitted signal becomes wider. A pulsed radar with pulsewidths sufficiently short to enable operation at a minimum range 528 onthe order of 0.25 meters fundamentally requires a frequency bandwidth inexcess of 200 MHz. Thus, the characteristics of the detection zone 500impact the system design in many ways. The detection algorithmprocessing that operates upon the received echoes, or signals, is alsoselected to operate with detection zones having particularcharacteristics. The detection algorithm is described below inconjunction with FIG. 12.

[0094] While a detection zone 500 to the side of the automobile isindicated, it should be recognized that other detection zones ofspecified sizes and shapes can be equivalently specified around theautomobile 508, such as those shown in FIG. 2. For example, frontalzones and rearward zones can be specified without departing from thisinvention. Typically, an automobile manufacturer specifies one or moredetection zones, including zone size, shape and position relative to theautomobile. The SOD system 504 can be configured so as to provide theappropriate side detection zone 500 regardless of where it is mounted onthe side of the automobile. Other probability zones 524 and otherminimum detection ranges 528 are also possible with this invention.

[0095] Also, while the SOD system 504 is shown at a position to the rearof the outside rear view mirror 532, the detection zone 500 and theresulting SLS sensor 504 location can be located anywhere along theperimeter of the automobile 508.

[0096] Referring now to FIGS. 11A and 11B, two different examples ofside detection zones 536, 548 are shown. In FIG. 11A, the maximumdetection zone boundary 536 is provided having a trapezoidal shape. Anexemplary SOD system provides eight azimuthal beams 540 a-540 h eachwith a different maximum detection range, as indicated in shading, andas determined by the detection algorithm that operates upon the beamechoes. The algorithmic control of the maximum detection range of eachof the eight beams defines the shape of an actual maximum detection zoneboundary 538 versus the specified maximum detection zone boundary 536.

[0097] The exemplary SOD system of FIGS. 11A, 11B has eight beams, eachwith a beam width of approximately fifteen degrees and with a totalazimuth scan of in excess of one hundred fifty degrees. It will berecognized by one of ordinary skill in the art that other numbers ofbeams (e.g. fewer than eight or more than eight) and scan angles arepossible without departing from the present invention. The particularnumber of antenna beams to use in a particular application is selectedin accordance with a variety of factors including but not limited to thedesired detection zone 500 and the width of the probability region 524.

[0098]FIG. 11B shows a detection zone 548 having a substantiallyrectangular shape of the maximum boundary. Again, an exemplary systemprovides eight azimuthal beams 552 a-552 h each with a different maximumdetection range as indicated in shading, the ranges 552 a-552 h beingdifferent from beams 540 a-540 h so as to form a different actualmaximum detection zone 550, indicated again by shading.

[0099] Referring now to FIG. 12, a flow diagram illustrates a process556 performed by the SOD processor, such as DSP 30 of FIG. 3. Theprocess contains major cycles and minor cycles. A minor cycle includesall detection operations that are performed utilizing a particular beamof the SOD antenna system. A major cycles includes operations that areperformed utilizing the full group of beams provided by the SOD antennasystem. In step 560 a major cycle is initiated and in step 564 a minorcycle is initiated. Initiating the major cycle 560 involves resettingthe beam to the first beam. Initiating the minor cycle 564 involvesresetting the beam to the next adjacent beam.

[0100] A chirp signal is initiated in step 570. During a chirp, a numbern of digital samples of the received signal are processed by the A/Dconverter 68 (FIG. 3), previously described, at a rate of f_(s) KHz, ina time period t. In the illustrative embodiment of FIG. 3, n=256,f_(s)=256 KHz, and t=1 msec.

[0101] While it will be recognized by those of ordinary skill in the artthat other numbers of samples at other rates are possible with thisinvention, selection of n and f_(s) determine the eventual frequency andrange resolution and thus must be selected to meet all system detectionrequirements.

[0102] In step 574, range thresholds are established at each beampointing angle in order to define the detection zone. Thus, at a givenbeam pointing angle, range detections that are either too far or toonear to the SOD system to fall within the detection zone are excluded byway of pre-established range thresholds. Echo magnitude windowthresholds are also established in step 574. Only object detections thatproduce echoes above a lower threshold and below an upper threshold,i.e. within the magnitude window, are considered valid. An echo that isbelow the lower threshold could be caused by system noise rather than anobject or target. An echo that is above the upper threshold could becaused by an interfering radar source, such as a radar transmission fromanother vehicle. Thus, both range and amplitude thresholds areestablished at step 574. It should be recognized that by altering therange thresholds, other specified detection zone shapes and sizes can beobtained.

[0103] In step 578, the data samples are truncated, whereby a smallnumber of samples at the beginning of the data set and at the end of thedata set are removed. The removed samples may contain unwanted artifactsdue to hardware limitations such as amplifier settling which can occurafter switching from one antenna beam to another, etc. In theillustrative embodiment, approximately two hundred fifty six datasamples remain after truncation.

[0104] Also in step 578, the remaining input samples are amplitudeweighted, or windowed. Recall from the discussion of FIG. 3 that thedata samples output of the A/D converter 68 are processed by an FFTwithin the DSP 30. Amplitude weighting of the data samples that areinput to an FFT can provide beneficial effects with regard to theminimum width of a spectral peak in the resulting frequency domain, andthe sidelobe magnitude associated with each FFT frequency bin.Essentially, with a spectrally pure frequency input to an FFT, inputweighting can make the resulting FFT output appear in a smaller numberof frequency bins than would occur with no input weighting. Since thefrequency difference between the received echo and the transmittedsignal relates directly to range, then the accuracy with which thereceived echo frequency is detected relates directly to range accuracy.Resolving a frequency to a single FFT bin or better is desirable.

[0105] In the exemplary SOD system a Chebyshev weighting function isapplied to the input samples. This weighting function provides a goodcompromise between spreading of a narrowband signal into multiple FFTbins and a good sidelobe suppression. It will also be recognized bythose of ordinary skill in the art that various amplitude weightingfunctions could be applied to the input data samples, and that theweighting may be applied with a variety of techniques including bothhardware and software multiplication's.

[0106] It will be recognized by those of ordinary skill in the art thatan FFT output is a frequency domain spectral representation of digitalinput samples to the FFT, where the digital input samples are samples ofa time domain analog signal. It will also be recognized that theparameters, including number of samples, n, and sample rate, f_(s),determine the frequency resolution, or bin width, f_(r), of the FFToutput by the relation f_(r)=f_(s)/n. Since the range of a target isassociated with the frequency of the received signal, the frequencyresolution correlates to a range resolution. As has been describedabove, a range resolution better than 0.6 meters is required in order toachieve the specified detection zone 500 (FIG. 10).

[0107] The exemplary SOD system provides two hundred fifty six datasamples, n, taken at a 256 KHz sample rate, f_(s). Thus, the resultingFFT frequency resolution, f_(s), is 1 KHz. The corresponding rangeresolution can be derived as follows. As mentioned earlier, in theillustrative SOD system 504 (FIG. 10), the frequency chirp is swept infrequency through 200 MHz in approximately 1 msec. One KHz thusrepresents a time period of approximately 5 nsec. In 5 nsec, radarenergy can propagate a distance of approximately 1.5 meters. Since theenergy propagates both to the target and back, the range to the targetrepresented by a 1 KHz signal from the A/D converter 68 (FIG. 3) isapproximately 0.75 meters. A 2 KHz signal represents a target at 1.5meters, etc. Thus, an FFT bin width of 1 KHz corresponds to a targetrange resolution of 0.75 meters. This measurement accuracy does not meetthe desired range resolution of less that 0.6 meters. Additionalprocessing is thus necessary.

[0108] In the exemplary embodiment at step 578, an zero paddingtechnique, recognized by those of ordinary skill in the art, is appliedto the input samples to the FFT in order to reduce the resulting FFT binwidth. It will also be recognized by those of ordinary skill in the artthat other techniques may be used to reduce the FFT bin width. Forexample, various interpolation techniques can be applied.

[0109] The windowed and zero padded input samples are operated on withan FFT operation in step 582. Time domain data samples are collectedfrom a chirp echo and a complex FFT is performed on the data samples.Subsequent processing can improve the measurement accuracy still furtheras described starting at process step 606.

[0110] In step 586, a magnitude calculation is performed on the complexFFT data, whereby the FFT real and imaginary parts of the complex FFTdata are used to calculate an FFT magnitude, hereafter called FFT outputdata, or an FFT output signal.

[0111] As the data samples associated with a particular beam steeringangle in azimuth can vary from one such data set to another, averagingof either time domain data samples or FFT output data associated with aparticular beam steer angle can reduce the rate of false detections, orimprove the false alarm rate. For example, where system noise or otherartifacts cause variation in the detected frequency and amplitude,averaging can reduce the impact of such artifacts and thus improvesystem performance.

[0112] In step 590, it is determined whether four chirps have beenprocessed, to be used in the aforementioned averaging. If four chirpshave not been processed, then processing returns to step 570 whereanother chirp is initiated. If four chirps have been processed, thenprocessing proceeds to step 602 in which the FFT output data from thefour chirps are averaged. Whereas the exemplary SOD system uses fouraveraged chirps for each beam within the minor cycle timing, it will berecognized by those of ordinary skill in the art that other numbers ofchirps can also be averaged.

[0113] In step 602, the resulting FFT output data is also rangenormalized. Whereas a radar return echo signal generally loses amplitudeproportional to range, and whereas increasing FFT bins are proportionalto increasing range, the FFT output data without range normalizationwould indicate an amplitude reduction across the FFT bins. Rangenormalization is provided to adjust the FFT bin amplitudes by scalefactors to minimize the amplitude range dependency of the FFT outputdata.

[0114] In step 606, the minimum and maximum range thresholds,frequencies R_(min) and T_(BeamRange) respectively, and minimummagnitude threshold, T_(object), are applied to the averaged FFT outputdata. The thresholds will be described more fully in conjunction withFIGS. 14A and 14B. Let it suffice to say here that the range thresholdslimit detections to those objects that are not too close and not too farfrom the SOD system, specific to the particular beam pointing angle, soas to detect objects as required within a specified detection zone, forexample detection zone 500 (FIG. 10).

[0115] Additional processing is required in order to achieve rangeaccuracy much better than the 0.6 meters as required by the exemplarydetection region 500. To this end, in step 606, first and secondderivatives of the FFT output data are computed by the DSP, as discussedfurther in conjunction with FIGS. 13A and 13B. As will be explainedbelow, range accuracy is significantly improved by use of the first andsecond derivatives of the FFT output data. In step 610, the negativegoing zero crossing of the first derivative and second derivative arecomputed as will be described.

[0116] The first and second derivatives of the FFT output data providean indication of whether a detection has or has not occurred. In thecase where a detection has occurred, the first and second derivativesalso provide a range to the detected object. Only those objects with arange within the range thresholds, and with sufficient magnitude areconsidered as valid targets by the SOD system in step 612, therebysetting a minor cycle detection flag. Step 612 concludes a minor cycleof processing in which detections associated with a single beam steeringangle are made.

[0117] In step 616, the minor cycle detection flag is stored in adetection table, or target detection report matrix. The table is shownin block 616 and includes columns corresponding to antenna beams androws corresponding to major cycles. Each entry in the table is a minorcycle detection flag. A “T” (true) indicates a minor cycle detection inthe associated beam detected during the minor cycle and F (false)indicates no minor cycle detection. Thus, the entry labeled 620,indicates detection in the first radar beam during a minor cycle. Acomplete row is obtained during each major cycle and each column of thedetection table 616 comprises successive minor cycle detection flags fora given beam steering angle.

[0118] After a minor cycle detection flag is stored in the table, it isdetermined in step 636 whether a major cycle has ended, i.e., whether aminor cycle detection flag has been provided for the last beam. If themajor cycle has not ended, then processing returns to step 564 on thenext beam and steps 564-616 are repeated for that beam. Alternatively,if the major cycle has ended, thus completing a row of the detectiontable, then the table is analyzed in step 644.

[0119] In step 644, minor cycle detection flags stored in the detectiontable are logically combined to provide an alert condition message witha reduced false alarm rate. An alert condition message 644 indicateseither the presence or absence of a target within the detection zone.For example, groups of four minor cycle detection flags such as thosegroups labeled 628 and 632 in the table may be logically combined toreduce the false alarm rate. The logically combined groups 628, 632 canspan both minor and major cycles, i.e. can span more than one column ormore than one row. In the illustrative embodiment, each group (e.g.group 628, 632) is a 2×2 matrix of detection flags. It will berecognized that group 632 is analyzed only at the completion of themajor cycle represented by the third row of the detection table, whereasgroup 628 is analyzed at the completion of the major cycle representedby the second row of the table.

[0120] Each combined group 628, 632 is processed to determine if atleast two detection flags within the group are true. In particular, twoor more true minor cycle detection flags in a 2×2 group of minor cycledetection flags yields an alert condition message that indicates atarget detection. Thus, processing of group 628 yields an alertcondition message resulting from consecutive detections in beam 1 duringconsecutive minor cycles. Processing of group 632 likewise yields analert condition message caused by detections in beams four and fiveduring one major cycle. Although a particular logical combination ofminor cycle detection flags has been described, it will be recognized bythose of ordinary skill in the art that other logical combinations ofminor cycle detection flags from one or multiple beams and from one ormore major cycles are possible with this invention. Upon completion ofmajor cycle processing in step 644, a new major cycle is initiated instep 560.

[0121] An alert condition message that indicates the presence of atarget in the detection zone can cause a system action appropriate forthe application. For example, a visual indication or audible alarm maybe provided to alert the driver that an object is within the detectionzone of a SOD system.

[0122]FIG. 13 shows minor cycle timing in relation to the chirp signaland processing steps. A minor cycle 648 is completed on each beamsteering angle in a predetermined period of time (e.g. 5.5 ms). Withinthat time period, four radar chirps 652 a-652 d are transmitted, thechirp echo returns are differenced from the transmission and thedifference is sampled during intervals 656 a-656 d, the differences areFFT analyzed during intervals 660 a-660 d, the FFT magnitudes arecalculated during intervals 664 a-664 d, and the four FFT output dataare averaged at interval 668. First and second derivatives are computedand analyzed during interval 672, and a minor cycle detection flag isgenerated at interval 676. The minor cycle 648 is then repeated for thenext beam. Note that the processing associated with intervals 668, 672,and 676 is performed coincidentally with sampling of the first chirp 680in the next minor cycle.

[0123] Randomly variable chirp offset intervals 680 a-680 d as describedabove in conjunction with FIGS. 5 and 5A provide a reduction in falsedetections caused by receptions from other radars including other SODsystems.

[0124] The overall minor and major cycle timing is selected in order todetect objects within a specified time period so that system actions canbe taken in an appropriate amount of time. For the illustrative SODsystem 504 (FIG. 10), an alert condition message is generated at thecompletion of each major cycle, i.e. at the completion of seven minorcycles. Thus, an alert condition message is generated in approximatelyonce every 50 msec. The CAN microprocessor 80 (FIG. 9) has anasynchronous update rate for the overall vehicle of approximately 300msec. Thus, several major cycles and associated alert condition messagesare generated during each CAN cycle. While specific timing selectionshave been described for the illustrative embodiment, it should berecognized that other timing selections are possible. For example,averaging of other than four FFTs, and use of other than seven beams arepossible. One of ordinary skill in the art, after reading thisdescription would clearly understand how to implement any necessarymodification to the timing structure of FIG. 13.

[0125] Referring now to FIGS. 14A and 14B an illustrative FFT outputdata curve 682 as may be generated in step 602 of FIG. 12 contains amagnitude detection peak 686 at a detection frequency 690 thatcorresponds to a particular detection range and one or more FFT bins. Itshould be recognized that the detection peak 686 from a single targetmay be broader than a single FFT bin. As mentioned above, the broadeningof the detected echo in the frequency domain is related to aspects ofthe particular FFT, in particular the weighting function that may beapplied the input data samples to the FFT. Broadening can also berelated to physical aspects of the echoing target, such as the dimensionof the target along the axis of the particular radar beam. An extendedtarget has an extended range and thus may appear as a broadenedfrequency corresponding to the range.

[0126] In general, the peak 686 of the FFT output data curve 682 for asingle target is a frequency 690 that corresponds to the range to thetarget. However, in association with FIGS. 15 and 16, it will be seenthat the peak can give a false impression of the range to the target insome cases. For now, let us assume that the peak of the FFT output datacurve indicates the range to the target.

[0127] As has been discussed in association with the detection algorithm556 of FIG. 12, the FFT frequency bins may not be sufficiently narrow toprovide the required accuracy required for a particular detection zone.Where the detected signal from a single target is spread between severalFFT bins, additional processing must be done on the FFT output data inorder to provide sufficient range resolution.

[0128] Consideration of the first derivative 694 of the FFT output datacurve 682 reveals that the peak 686 of the FFT output data curve 682 iscoincident with the negative going zero crossing 698. Thus, the negativegoing zero crossing 698 of the first derivative 694, provides thefrequency 690 of the FFT peak 686, and thus, the corresponding range tothe target. Range detection in this manner provides greater accuracythan simply determining the range by determining the fractionalfrequency bin of the FFT output data curve 682 in which the peak occurssince the peak may occur in multiple FFT bins.

[0129] It should be recognized that although FIGS. 14A and 14B show theFFT output data curve 682 and FFT first derivative 694 as smoothcontinuous waveforms, the FFT output data, represented by FFT outputdata curve 682, and first derivative 694 are comprised of discontinuousdigital samples. As a result, finding the zero crossing of the firstderivative 694 requires additional processing. If the spacing of thepoints of the first derivative 694 are sufficiently close, then adjacentfirst derivative data points lie approximately on a straight line. Withthis approximation, the zero crossing 698 can be found by a conventionalsimilar triangles technique. In the illustrative embodiment, spacingbetween data points on the order of 500 Hz has been found to besufficient. However, it will be recognized by those of ordinary skill inthe art that other data point frequency spacings can be used with thisinvention. It should be noted that the zero crossing 698 of the firstderivative so determined can be at any frequency and need not be at thecenter of an FFT bin, thus the resolution in frequency and associatedrange resolution is greatly improved as compared to the conventionaltechnique of determining the frequency bin in which the FFT peak 686occurs.

[0130] For complex echo returns, for example from multiple targets orfrom a target that is extended along the beam axis, the negative goingzero crossing 702 of the second derivative 706 is determined andprovides greater range discrimination that the use of the firstderivative in the above manner. The negative zero crossing 702 of thesecond derivative 706 of the FFT output data curve 682 corresponds to anegative inflection point 710 of the FFT output data curve 682. Anegative inflection point 710, is a point where the slope of the FFToutput data curve 682 changes from curving upward to curving downward.

[0131] It will be recognized that the rising portion of an FFT outputdata feature, for example 714, has a discrete negative inflection point710, whereas the peak 686 of the feature can be extended along thefrequency axis. Although the broadening of an FFT output data feature isrelated to both to mathematical characteristics of the FFT, such as theinput sample weighting, and to dimensional characteristics of thetarget, the shape of the rising portion of the FFT output data feature714 is, to a first order, related only to the FFT mathematicalcharacteristics. The negative inflection point 710 on the rising portionof the FFT output data feature 714 can be used to predict the frequencyof the FFT peak 686 as may be desirable where the peak 686 is nototherwise distinct. In essence, for a relatively broad range of targetdimensional characteristics, the distance 716 between the negativeinflection point 710 and the FFT peak 686, is known and constant,determined only by the mathematics of the FFT. Thus, by finding thefirst negative inflection point 710 of the rising portion of an FFToutput data feature 714, and by using the negative inflection point 710to predict the position of the peak 686, the range to the target can befound even for more complex FFT output data curve spectral shapes.

[0132] As noted in conjunction with step 606 in FIG. 12, thresholdsR_(min) 717 and T_(BeamRange) 718, specific to the particular beampointing angle, are applied to the calculated frequency of an objectrange as calculated by either first derivative or second derivativetest. In particular, those peaks within the thresholds will beconsidered valid. Similarly, only those peaks that are above a magnitudethreshold T_(object) 719 are considered valid.

[0133] Referring now to FIG. 15 an example curve 728 corresponding toFFT output data as may be caused by a target that is extended along thebeam axis so as to result in a nearly flat top FFT output data curve forwhich a peak would be indeterminate or result in multiple peakdetections 724, 732 is shown. Also shown is curve 736 which correspondsto FFT output data resulting from a non-extended target. Comparison ofthe FFT output data curve 728 with the FFT output data curve 736 revealscoincident negative inflection points 720, 740. Thus, computation of thesecond derivative of the FFT output data curve 728 and use of theresulting negative zero crossing to determine peak 724 by adding theknown distance, i.e. frequency, between negative inflection point 720and peak 724 as described above, results in an accurate rangedetermination.

[0134] Referring now to FIG. 16, another curve 744 corresponding to anexample of FFT output data as may be caused by simultaneous receipt ofechoes from two targets, of which one is weaker than the other is shown.Note that a strong echo can cause the curve portion 748 of the compositeFFT output data curve 744 associated with the weaker echo to have nopeak at all, while the curve portion 752 from the stronger target has adistinct peak 756. Still, the negative inflection points 760, 764 foundby computing the second derivative of the composite FFT output datacurve 744 can be used to predict the range of both targets. Detection ofthe two negative zero crossings of the second derivative of FFT outputdata curve 744 and use of the known distance, like distance 716 of FIG.14A, results in detection of peak frequency points 768 and 756. A firstnegative inflection point 760 is used to find the peak 768 correspondingto the first target range, though a peak 768 does not in fact exist. Asecond negative inflection point 764 is used to similarly find the peak756 corresponding to the second target range. Since the frequency peakscorrespond to target range, the range to both the strong and weak targetcan be discriminated. Thus, use of the second derivative of the FFToutput data provides range discrimination for targets represented bymore complex FFT output data curves.

[0135] Having described the preferred embodiments of the invention, itwill now become apparent to one of ordinary skill in the art that otherembodiments incorporating their concepts may be used.

[0136] It will be appreciated by those of ordinary skill in the art thatthe particular boundaries between portions of the radar system 10 can bevaried from that described herein above. As examples, the receiver 24may include parts of the electronic control circuits 34 or parts of thereceiver, such as an A/D converter (FIG. 2), may be provided in theelectronics portion 28 of the system. Depending upon the selectedimplementation of the various components, one or more portions of theradar system may be integrated onto one or more hybrid circuits,modules, or subassemblies.

[0137] It is felt therefore that these embodiments should not be limitedto disclosed embodiments but rather should be limited only by the spiritand scope of the appended claims. All publications and references citedherein are expressly incorporated herein by reference in their entirety.

What is claimed is:
 1. A radar transmitter comprising: a DSP having anoutput terminal at which a digital output word is provided; a D/Aconverter having an input terminal coupled to the output terminal of theDSP and an output terminal at which an analog ramp signal is provided;and a VCO having an input terminal responsive to the analog ramp signaland an output terminal at which a frequency modulated signal isprovided.
 2. The radar transmitter of claim 1 further comprising asmoothing circuit having an input terminal coupled to the outputterminal of the D/A converter and an output terminal coupled to theinput terminal of the VCO.
 3. The radar transmitter of claim 1 furthercomprising an up-converter having an input terminal coupled to theoutput terminal of the VCO and an output terminal at which a transmitsignal is provided.
 4. The radar transmitter of claim 3 wherein saidup-converter comprises a DRO.
 5. The radar transmitter of claim 1wherein said VCO comprises: an amplifier having an input terminal towhich a terminator is coupled and an output terminal; a phase shifterhaving an input terminal coupled to the output terminal of theamplifier, an output terminal at which a frequency modulated signal isprovided, and a control terminal; and a DR positioned to couple energyfrom the output terminal of the phase shifter to the input terminal ofthe amplifier.
 6. The radar transmitter of claim 1 further comprising: amemory in which is stored a look-up table containing a plurality of DSPoutput words in association with an expected transmit frequency; afrequency detection circuit for detecting the frequency of a transmitsignal generated by the frequency modulated signal; and a comparator forcomparing a DSP output word from which the detected frequency isgenerated with a DSP output word stored in the look-up table inassociation with the detected frequency to provide an error value. 7.The radar transmitter of claim 6 wherein said frequency detectioncircuit comprises a DR having a fundamental frequency substantiallyequal to the detected frequency.
 8. The radar transmitter of claim 6further comprising an offset generator for adjusting the output word ofthe DSP in response to the error value.
 9. The radar transmitter ofclaim 1 further comprising a random number generator for varying atleast one parameter of the analog ramp signal.
 10. The radar transmitterof claim 9 wherein the at least one parameter is selected from an offsetinterval and a voltage range.
 11. The radar transmitter of claim 1wherein said transmitter is mounted on a vehicle and comprises a sideobject detection module of a near object detection system.
 12. A methodfor compensating for temperature induced variations in a radar systemcomprising the steps of: generating, from a predetermined sequence ofdigital words, a transmit signal having a frequency associated with arespective word of the predetermined sequence of digital words; storingeach word of the predetermined sequence of digital words in associationwith an expected transmit signal frequency; detecting the actualfrequency of the transmit signal; comparing the digital word from whichthe actual frequency is generated to a digital word stored inassociation with the actual frequency to provide an error value; andadjusting each word of the predetermined sequence of digital words inresponse to the error value.
 13. The method of claim 12 wherein saidadjusting step includes offsetting each word of the predeterminedsequence of digital words by the error value.
 14. The method of claim 12wherein said detecting step includes detecting the actual frequency ofthe transmit signal with a DR having a fundamental frequencysubstantially equal to the actual frequency.
 15. A method for reducinginterference between radar systems comprising the steps of: generating aramp signal for controlling to a VCO; and randomly varying at least oneparameter of said ramp signal.
 16. The method of claim 15 wherein saidramp signal comprises a plurality of cycles, each comprising an offsetportion, a ramp portion, and a CW portion and wherein said at least oneparameter is randomly varied in at least one of said plurality ofcycles.
 17. The method of claim 16 wherein the at least one parameter isselected from the duration of said offset portion and a voltage range ofsaid ramp portion.
 18. A method for compensating for non-linearoperation of a VCO in a radar transmitter, comprising the steps of:measuring the output frequency of the VCO in response to a plurality ofinput voltage levels to the VCO to provide a characteristic curve of theVCO; determining a complementary waveform with respect to thecharacteristic curve; and providing a sequence of output words of a DSPoperable to generate a transmit signal for generating the complementarywaveform.